HAL Id: hal-02087695
https://hal.univ-brest.fr/hal-02087695
Submitted on 2 Apr 2019
HAL is a multi-disciplinary open access
archive for the deposit and dissemination of sci-
entic research documents, whether they are pub-
lished or not. The documents may come from
teaching and research institutions in France or
abroad, or from public or private research centers.
L’archive ouverte pluridisciplinaire HAL, est
destinée au dépôt et à la diusion de documents
scientiques de niveau recherche, publiés ou non,
émanant des établissements d’enseignement et de
recherche français ou étrangers, des laboratoires
publics ou privés.
A Cost-Eective Wideband Switched Beam Antenna
System for a Small Cell Base Station
Petros Bantavis, Christos I Kolitsidas, Tzihat Empliouk, Marc Le Roy, B.
Lars G. Jonsson, George. A Kyriacou
To cite this version:
Petros Bantavis, Christos I Kolitsidas, Tzihat Empliouk, Marc Le Roy, B. Lars G. Jonsson, et al..
A Cost-Eective Wideband Switched Beam Antenna System for a Small Cell Base Station. IEEE
Transactions on Antennas and Propagation, Institute of Electrical and Electronics Engineers, 2018,
66 (12), pp.1-4. �10.1109/SaPIW.2018.8401663�. �hal-02087695�
0018-926X (c) 2018 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TAP.2018.2874494, IEEE
Transactions on Antennas and Propagation
IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. , NO. , 2018 1
A Cost-effective Wideband Switched Beam Antenna
System for a Small Cell Base Station
Petros I. Bantavis, Christos I. Kolitsidas, Member, IEEE, Tzihat Empliouk, Marc Le Roy, Member, IEEE,
B. L. G. Jonsson and George A. Kyriacou Senior Member, IEEE
Abstract—A wideband switched beam antenna array system
operating from 2 to 5 GHz is presented. It is comprised of
a 4 × 1 Vivaldi antenna elements and a 4 × 4 Butler matrix
beamformer driven by a digitally controlled DP4T RF switch. The
Butler matrix is implemented on a multilayer structure, using
90
◦
hybrid couplers and 45
◦
phase shifters. For the design of
the coupler and phase shifter we propose a unified methodology
applied, but not limited, to elliptically shaped geometries. The
multilayer realization enables us to avoid microstrip crossing
and supports wideband operation of the beamforming network.
To realize the Butler matrix we introduce a step by step
and stage by stage design methodology that enables accurate
balance of the output weights at the antenna ports to achieve
stable beamforming performance. In this work we use a Vivaldi
antenna element in a linear four element array, since such
element supports wideband and wide-scan angle operation. A
soft condition in the form of corrugations is implemented around
the periphery of the array, in order to reduce the edge effects.
This technique improved the gain, the side lobes and helped
to obtain back radiation suppression. Finally, impedance loading
was also utilized in the two edge elements of the array to improve
the active impedance. The proposed system of the Butler matrix
in conjunction with the constructed array can be utilized as a
common RF front end in a wideband air interface for a small
cell 5G application and beyond as it is capable to simultaneously
cover all the commercial bands from 2 to 5 GHz.
Index Terms—Wideband, Butler matrix, RF front end, 5G,
Vivaldi array, soft surface, edge impedance loading, small cell.
I. INTRODUCTION
M
ODERN wireless communications are driven by high
quality end user experience providing high data rates,
low latency and extended coverage. It is predicted that with the
massive deployment of the Internet of things (IoT), the number
of connected devices will increase exponentially. Network
densification, [1], conjointly with heterogeneous networks
(HetNets) will be in the heart of future wireless networks offer-
ing increased network capacity and spectral aggregation, [2].
Additionally, exploiting spatial multiplexing we can further
increase the network capacity and offer higher signal-to-noise
Petros. I. Bantavis was with the Lab-STICC at the university ENSTA-
Bretagne in Brest, France. Now he is with institut d’ electronique
et de Telecommunications de Rennes, Universite de Rennes 1, e-mail
(petros.bantavis@univ-rennes1.fr).
Christos I. Kolitsidas and B. L. G. Jonsson are with with the Department of
Electromagnetic Engineering, School of Electrical Engineering, KTH Royal
Institute of Technology, SE-100 44, Sweden. e-mail: (chko@kth.se).
Tzihat Empliouk and Georgios. A. Kyriacou are with the Microwave Lab-
oratory, School of Electrical Engineering, Democritus University of Thrace,
Xanthi, Greece. e-mail: (gkyriac@ee.duth.gr).
Marc Le Roy is with Lab-STICC at Universit
´
e de Bretagne Occidentale
(UBO) in Brest, France. e-mail: (marc.leroy@univ-brest.fr)
controller
NETWORK 4x4
ARRAY 4x1
Fig. 1: Illustration of dividing data traffic into macro and small
cells and the block diagram of the proposed system.
ratio (SNR) by focusing the RF power to the desired direction.
Small base stations, [3], are in the center of the emerging next
generation wireless networks playing an instrumental role to
the aforementioned demands. Small cell base stations due to
their foreseen massive deployment should also be very cost
effective and budget friendly devices.
Recent works, [4], [5], have shown the potential of switched
beam systems based on a Butler matrix, [6]. These works
have been developed for the 60 GHz ISM band that can
accommodate high data rates but with increased path loss. The
resulted beamforming in the 60 GHz band will be utilized to
electrically optimize the line of sight link. In [7], a 2 × 2
circularly polarized array with a Butler matrix capable to
operate in about 48% bandwidth was presented. Slomian et.al.
in [8] provided an array configuration with a Butler matrix
capable for dual linear and dual circular polarization operating
in 5.2 - 5.4 GHz. In both approaches the bandwidth was
limited in less than 50% with severe impact on the second
approach. A narrow band Butler matrix integrated with a patch
antenna array is presented in [9]. In [10], wider bandwidth was
achieved, however only simulated results are presented.
A wideband switched beam antenna system that will act as
an RF air interface and will be software defined is proposed.
Either a universal software radio peripheral (USRP) or a
micro-controller (µC) can be used to control electronically
the beamforming network. The USRP is proposed to evaluate
the whole system through lab experiments, however a micro-
controller is the best candidate for a massive implementation.
Integrating an analogue RF switched beam system with a
micro-controller offers the advantage of low cost and com-
plexity with spatial multiplexing capabilities. The proposed
0018-926X (c) 2018 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TAP.2018.2874494, IEEE
Transactions on Antennas and Propagation
2 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. , NO. , 2018
system presented in Fig. 2 is comprised of a 4 × 1 Vivaldi
linear array fed by 4 × 4 Butler matrix connected to Double
Pole Four Throw (DP4T) switch. The system will be controlled
by a µC and have a single transmit/receive chain. This set up
can offer full duplex capabilities. In this way the analogue
beamforming of the Butler matrix and the digital capabilities
of the µC are exploited resulting in a hybrid system. Since
only one µC is required for the proposed system with only
one full duplex channel, the cost is significantly reduced. The
system is operating from 2 to 5 GHz providing more than one
octave of usable bandwidth.
Wideband beam forming networks have already been pro-
posed in literature as a base of switched beam systems and
they have been implemented as single layer or multilayer
structures. Single layer wideband Butler matrix structures have
been introduced in [11], [12] where tapered line directional
couplers and Schiffman phase shifters were used to implement
the Butler matrix. Wide bandwidth was achieved but with
the penalty of larger physical size. Several implementations
of Butler matrices have adopted the multilayer slot coupled
technology which was originally proposed in [13], [14] for the
implementation of the required 90
◦
hybrids and phase shifters.
In [15], [16] this concept was extended into elliptically shaped
geometrical structure, whereas in [17] a variation of the
rectangular aperture slot coupler was presented that eliminated
the magnitude rippling on the transmission coefficient. A
hexagonal shaped approach was adopted in [18] and a wide-
band Butler matrix was designed, implemented and measured.
In [19], a trapezoidal shaped geometry was introduced for
the implementation of a wideband Butler matrix and only
simulated results were presented for the corresponding beam
performance. The last two approaches used more a brute
force technique to select the corresponding geometries and no
theoretical motivation or explanation was adequately provided
for the design procedure of the different geometrical shapes.
Fig. 2: The hybrid switched beam system block diagram.
In our previous works, [20], [21], we have focused on
developing switched beam system for Ultra Wide Band (UWB)
applications. In [22] we have presented an early version of the
proposed system. Lower frequency bands and especially the
sub 6 GHz communication bands have received little to no
attention for small base stations as this requires challenging
wideband or multiband RF circuitry development and large
dimensions due to the corresponding wavelength. Due to the
required backwards compatibility with the current wireless
communication systems increased attention is expected to the
sub 6 GHz frequencies for small cell base stations. Next
generation wireless communication systems are expected to
extend the capabilities but continue the support of the current
TABLE I: Ideal complex excitation currents at the antenna
array elements.
BEAMS ANT1 ANT2 ANT3 ANT4
1L 1∠0
◦
1∠135
◦
1∠ − 90
◦
1∠45
◦
2L 1∠0
◦
1∠45
◦
1∠90
◦
1∠135
◦
2R 1∠0
◦
1∠ − 45
◦
1∠ − 90
◦
1∠ − 135
◦
1R 1∠0
◦
1∠ − 135
◦
1∠90
◦
1∠ − 45
◦
systems. An illustration of the application of small cell base
stations can be depicted as in Fig. 1, where small cells can be
deployed on buildings or high traffic areas.
The originality of this work refers to the design, fabrication
and testing of an integrated wideband switched beam phased
array system. We present a system in the sub 6 GHz band with
a 4×4 Butler matrix and a 4×1 electrically connected Vivaldi
array. In section II, we introduce the design methodology and
we propose the unified theoretical approach on the design
of multilayer 90
◦
wideband hybrids and phase shifters. This
approach was applied in the elliptically shaped geometry but
it can easily be extended in any other geometrical shape.
The overall Butler matrix design is also presented in this
section. The proposed Butler matrix is connected with a
linear 4 × 1 linear Vivaldi array that is introduced in section
III. The combination of edge impedance loading and a soft
condition at the array’s periphery are utilized to achieve the
wideband characteristics of such a small array. The overall
system performance is presented in section IV. Finally, the
conclusions of this work are summarized in section V.
II. DESIGN METHODOLOGY
A. 4 × 4 Butler matrix design
Butler matrix is one of the most widespread analogue
beamforming networks. It is composed of N inputs (excitation
ports) and N outputs (antennas ports) where N = 4. The
number of couplers is equal to (N/2) log
2
(N) and the number
of phase shifters to (N/2)[log
2
(N) − 1], [6]. Exciting one of
its N inputs produces uniform amplitudes at the output ports
with a phase difference ∆φ. This phase difference between
the output ports is different for every input port excitation and
at the same time it is the factor that steers the beam in the
desired direction in space. As a result of the uniform excitation
currents the radiation field is of the form sin x/x. Exciting
the input ports of a N = 4 matrix separately, four orthogonal
beams are produced, so there is no significant interference
between the consecutive beams. The operation of the Butler
matrix and the resulting ideal excitation power vectors can be
summarized in TABLE I.
In order to induce wideband characteristics the present But-
ler matrix consists of multilayer couplers and phase shifters.
Another advantage of the multilayer technology is that it
avoids undesired line crossings. Thus, a compact and wideband
beamforming network in the sub 6 GHz communication band
from 2 to 5 GHz as depicted in Fig. 2 is designed, fabricated
and tested. In Fig. 3(a) the schematic of a 4 × 4 Butler
matrix is depicted and it consists of four 90
◦
hybrid couplers
and two 45
◦
phase shifters. This approach is extended for
both the hybrid couplers and phase shifters, especially when
0018-926X (c) 2018 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TAP.2018.2874494, IEEE
Transactions on Antennas and Propagation
BANTAVIS et al.: A WIDEBAND SWITCHED BEAM SYSTEM FOR SMALL CELL APPS. 3
implemented with elliptical patches and slots. This multilayer
technology besides the wideband operation overcomes the
usual problem of interconnecting line crossings encountered
in printed microstrip Butler matrices. An indicative layout
geometry of the technology is illustrated in Fig. 3(b) where
the continuous and dashed lines represent different layers. It is
clear that no line crossing occurs in the same layer. In Fig. 3(b)
it is shown the corresponding physical implementation. In Fig.
3(c) the multilayer configuration of the coupler is depicted,
where two elliptical patches are printed on the top and bottom
surfaces of two grounded substrates pressed back-to-back at
their common ground plane. An elliptical slot is etched on
their common ground plane which is oriented orthogonally
to the elliptical patches. To build the respective prototype the
top and bottom substrates are fabricated separately and then
pressed and glued together.
(a)
1L
2R 2L 1R
ANT2
ANT1
ANT3
ANT4
dw
ds
dl
dps
dpl
dpw
(b)
ds
dl
dw
(c)
Fig. 3: (a) Butler matrix block diagram with the continuous
and dashed lines defining different layers (b) Top layered view
of the developed Butler matrix where the middle copper layer
is with brown color and the top bottom copper layers with
light dark and light yellow shades. The denoted dimensions in
mm: d
s
= 13.7, d
l
= 14.9, d
w
= 6.1, d
ps
= 13, d
pl
= 13.8,
d
pw
= 5.9. (c) 3D layered illustration of the hybrid coupler
and top perspective same as (b).
The final layout implementation is shown in Fig. 3(b) where
the three distinctive layers are visible. The dimensions of the
coupler and phase shifter that are integrated in the Butler
matrix are shown in Fig. 3(b). A step by step and a stage
by stage design procedure is proposed for the Butler matrix
design. The design is divided in two steps and three stages
as indicated in Fig. 3(b). The first step is to design the
individual subnetworks, which are the hybrid coupler and the
phase shifter of the Butler matrix. After the successful design
of the two individual components the coupler is connected
with the phase shifter resulting in a two port network. At
this stage the components are numerically optimized again to
achieve optimal cooperative performance. Next, at stage 2 the
Butler matrix is designed. The magnitude and phase behavior
of this network are evaluated and numerically optimized.
The design procedure continues with the second step, (see
Fig. 3(b)), where the equiphased transmission line network is
designed separately. This line network will align and order in
sequence the Butler matrix output ports to correspond to the
linear antenna array ports enabling their integration. Finally,
in the third stage the circuitry from the two previous stages
is integrated and the overall performance of the final Butler
matrix is evaluated.
B. Subnetworks
The adopted technology for the hybrid couplers and phase
shifters consists of three conductive layers interleaved with two
dielectric layers as shown in Fig. 3(b). The top and bottom
layers are coupled through an aperture slot that is located
in the middle layer. The coupling phase and bandwidth are
controlled by the shape of the two conductive patches and
the slot. Rogers 4003C with thickness h=0.813 mm is used
for a substrate as a design material that provides a good
compromise between losses and cost for RF circuit. The phase
shifters adopted in this work are based on the same principle as
the hybrid couplers. However, they provide a constant phase-
shift versus frequency with respect to a corresponding uniform
transmission line.
In this work we propose a unified design methodology for
the 90
◦
hybrid coupler and phase shifter based on [15], [16]
which in turn are based on the original work from [13] since
the designs share a similar geometrical cross section. The
coupling coefficient can be calculated as:
C =
Z
e
− Z
o
Z
e
+ Z
o
(1)
where Z
e
and Z
o
denote the even and odd mode character-
istic impedances. The feeding microstrip line characteristic
impedance is calculated using the geometrical mean Z
0
=
√
Z
e
Z
o
. The impedances Z
e
, Z
o
are derived from [13] using
the image theory as:
Z
e
=
60π
√
r
K(k
1
)
K
0
(k
1
)
(2)
Z
o
=
60π
√
r
K
0
(k
2
)
K(k
2
)
(3)
where K(k) is the elliptical integral of first kind and K
0
(k)
its complementary given by K
0
(k) = K(
√
1 − k
2
). This ratio
of elliptic integrals can be approximated either numerically or
using the formulas given in [13]. The elliptic modulus k
1
, k
2
are given based on the geometrical characteristics as follows:
k
1
=
s
sinh
2
(π
2
E
s
/(16h))
sinh
2
(π
2
E
s
/(16h)) + cosh
2
(π
2
E
w
/(16h))
(4)
k
2
= tanh(π
2
E
w
/(16h)) (5)
where E
s
∈ {d
s
, d
ps
} and E
w
∈ {d
w
, d
pw
} with respect to
the notation of Fig. 3(b) and Fig. 3(c). A π/4 multiplier has
0018-926X (c) 2018 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TAP.2018.2874494, IEEE
Transactions on Antennas and Propagation
4 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. , NO. , 2018
been added when compared to the original formulas from [23].
This multiplier indicates the area ratio between a rectangle,
A
rect
, with sides (a, b) and the corresponding inscribed ellipse
with area A
ell
, as A
ell/A
rect
= (πab/4)/(ab) = π/4. This
π/4 multiplier accounts for the ellipticity of the adopted
geometry. An additional step is required for the phase shifter.
Its corresponding phase shift when referenced to a microstrip
line is given in [16]:
∆φ = π/2−arctan
"
sin(β
eff
d
pl
)
√
1 − C
2
cos(β
eff
d
pl
)
#
+β
ms
l
ms
(6)
where β
eff
= β
0
√
r
, β
ms
is the corresponding microstrip
propagation constant, d
pl
= λ
ms
/4 = d
l
with λ
ms
the
effective microstrip wavelength and l
ms
the microsrip line
reference length. Equation (6) reveals that there are two
degrees of freedom to achieve the desired phase shift, the
coupling coefficient and the l
ms
. Hence in this particular case
the coupling coefficient C of the phase shifter is chosen based
on the corresponding length required from the layout. This
concludes the proposed unified approach for the design of the
elliptically shaped aperture coupled phased shifter and hybrid
coupler. Our proposed unified design methodology is not lim-
ited to elliptically shaped coupling geometries and can easily
be extended to other coupling geometries simply by modifying
accordingly the area ratio. Calculating such multiplier for any
other geometry such as diamond shaped or hexagonal can
provide a priori analytical estimates for the corresponding
physical dimensions. The wideband characteristics and the
small physical footprint of the coupler and the phase shifter are
attributed to the folded aperture coupling occurring in different
layers.
1) Prototypes Construction: For the construction process
silver epoxy was used in order to connect the two printed
circuit boards (PCB) layers. The ground metalization is re-
tained in both PCB layers to avoid via padding for the coaxial
connector and manufacturing ease. Only board routing is
required with this methodology. The first step was to spread
this epoxy over the whole surface of one layer and then press
together the metalized layers (ground planes) mechanically.
During the pressing, the PCB is heated for 10 minutes at 100
◦
C. This heating helps the epoxy to get similar conductivity
as the copper and the layers adhere effectively without air-
bubbles trapped in between.
2) Hybrid Coupler: The first constructed subnetwork is the
90
◦
hybrid coupler obtained from the overall design process
described above. Since the integrated design is evaluated the
deviation from the ideal results will indicate the difference
resulted from the interconnection of individually designed
components. Starting the equations (1)-(5) the initial ellipses
dimensions denoted in Fig. 3(b) (values in mm) d
s
= 8.1,
d
l
= 12.99 and d
w
= 6.1 are obtained. Ideally, exciting
port 1, as designated in the inset of Fig. 4 the coupler
equally divides the power into ports 2 and 3 with a 90
◦
phase difference, while port 4 remains isolated. The simulation
and measurement results for the magnitude of the constructed
coupler are depicted in Fig. 4. An acceptable variation of
±0.75 dB from the ideal value of −3 dB is observed in the
2-5 GHz band of interest, whereas the reflection coefficient
and the isolation are kept below -18 dB.
3) Phase Shifter: The other internal component of the
Butler matrix is the phase shifter. A 4 × 4 Butler requires
two phase shifters with ∆φ = 45
◦
as illustrated in Fig. 3(a).
A differential phase shifter is utilized which is based on the
same multilayer technology by open ending and removing the
transmission lines from ports 3 and 4 of the hybrid coupler.
Using equations (1) - (6) we obtain the initial values in mm
d
ps
= 8.92, d
pl
= 12.99 and d
pw
= 6.22. This phase shifter
is able to provide constant phase shift when referenced to a
corresponding transmission line. The topology of the Butler
matrix has this property inherently since a transmission line
is required to connect the two remaining ports (see Fig. 3(a))
of the first stage of hybrid couplers. These initial dimensions
are optimized to achieve the required constant phase shift
throughout the bandwidth of the operation. The constructed
phase shifter and the referenced line are depicted in the inset
of of Fig. 5. Excellent agreement between simulation and
measurements of the designed subnetwork is illustrated in Fig.
5. Very low losses (< 1 dB) are also observed in the band of
interest.
In Fig. 6 the phase responses for both the coupler and
the phase shifter are illustrated. For the coupler the phase
difference between the output ports 2, 3 when port 1 is excited
is illustrated (inset of Fig. 4), whereas for the phase shifter
the achieved phase difference with respect of the reference
transmission line ∆φ = ∠S
21
− ∠S
34
is depicted in Fig.
5. Very good agreement between the simulation and the
measurements and the ideal value is observed for the sub-
band 2.5-4 GHz. For the low and the high end of the band,
higher but acceptable deviations up to 5
◦
are observed. The
deviations can also be traced in the manufacturing process
of the subnetworks where we utilized a milling process for
this implementation. The milling process will not protect the
copper of the PCB of being contaminated with particles that
could potentially result in higher resistivity of the ground
connection. The milling process can also affect the thickness
of the substrate which can impact the phase. The final Butler
matrix is constructed using etching process and a semi-clean
room environment to avoid the aforementioned issues.
C. Butler Matrix Network Implementation and Measurements
After the implementation and verification of the two key
components of the Butler matrix, the procedure continues with
the implemented network verification. An ideal Butler matrix
will equally divide the power from an input port and depending
on the activated port, four different sets of phase sequences
with ∆φ = ±135
◦
and ±45
◦
are obtained corresponding to
four respective beams.
The final constructed Butler matrix is depicted in Fig. 7
and has overall dimensions 150 mm× 110 mm. Emphasis has
been given to the outputs of the Butler matrix to be placed in
sequential order and thus be ready to connect with the antenna
array. Furthermore, the distance between the output ports is set
to be same as for the designed linear Vivaldi array that will be
described in section III. The latter step has been taken so that