For Review Only
Abstract—In this paper, a novel dual-band dual-polarized
(DBDP) array antenna with low frequency ratio and integrated
filtering characteristics is proposed. By employing a dual-mode
stub-loaded resonator (SLR) to feed and tune with two patches,
the two feed networks for each polarization can be combined,
resulting in the reduction of the feed networks and the input ports.
In addition, owing to the native dual resonant features of the SLR,
the proposed antenna exhibits 2
nd
-order filtering characteristics
with improved bandwidth and out-of-band rejections. The
antenna is synthesized and the design methodology is explained.
The coupling coefficients between the SLR and the patches are
investigated. To verify the design concept, a C/X-band element
and a 2 × 2 array are optimized and prototyped. Measured results
agree well with the simulations, showing good performance in
terms of bandwidth, filtering, harmonic suppression and
radiation at both bands. Such an integrated array design can be
used to simplify the feed of a reflector antenna. To prove the
concept, a paraboloid reflector fed by the proposed array is
conceived and measured directivities of 24.6 dBi (24.7 dBi) and
28.6 dBi (29.2 dBi) for the X-polarization (Y-polarization) are
obtained for the low- and high-band operations, respectively.
Index Terms—Antenna array, dual-band, dual-polarized,
filtering, integrated, stub-loaded resonator.
I. INTRODUCTION
IRELESS terrestrial and space/air-borne applications
have an increase demand for the antenna with the
features of dual-band dual polarizations [1]. Polarization
diversity is usually desirable for enhancing the information
content and combating the multi-path fading [2]-[3]. Dual-band
or multi-band operations, on the other hand, could increase the
versatility of the systems. For multifunction radar applications,
different frequency operations are also required to share the
same aperture in order to reduce the cost and weight of the RF
frontend [4]-[6].
Manuscript submitted on June 29, 2016; This work is supported by the
project “DIFFERENT” funded by EC FP7 (grant no. 6069923). YW is
supported by UK EPSRC under Contract EP/M013529/1.
C. X. Mao, S. Gao and Q. Luo are with the School of Engineering and
Digital Arts, University of Kent, UK. (e-mail: cm688@kent.ac.uk, s.
gao@kent.ac.uk).
Y. Wang is with the Department of Engineering Science, University of
Greenwich, UK.
Q. X. Chu is with the School of Electronic and Information Engineering,
South China University of Technology, China.
Traditionally, dual-band operations of an antenna can be
implemented by using two single-band elements or using one
dual-band element. Employment of two single-band elements
is always regarded more flexible for designing the dual-band
antennas with high frequency ratios. In [4]-[8], perforated
patches are used to design the dual-band arrays with dual
polarizations characteristics while sharing the same aperture.
In [9]-[13], the radiating elements with different resonant
frequencies are interlaced for dual-band operations. One of the
challenges in these array designs is the congestion of the
feeding networks since each band and each polarization is
usually excited separately. This would be much challenging as
the frequency ratio decreases (much smaller than 2). To the
best of the authors’ knowledge, very few work has been
reported focusing on the DBDP array with low frequency ratio.
To implement a DBDP array with a low frequency ratio, the
dual-band element is believed to be more suitable. The main
challenge of using a dual-band element is how to layout of the
radiating elements to avoid grating lobes. Other challenges
include complex feeding structures and mutual coupling
between the two operation bands. A kind of combined feed is
proposed to reduce the number of input ports [6]-[8]. However,
these essentially involve separate feeding networks for each
band then combined using a “duplexing” device with the
assistance of the band-notch structures and matching stubs. The
number of the feeding networks as well as the congestion are
not reduced. Thus, they are not very suitable for solving the
problems in low frequency ratio DBDP antenna design.
To overcome the problems, an integrated design concept is
proposed in this paper. The integration of filter and antenna
has been intensively investigated during the past several years
[14]-[20]. In [20], an approach of using a multi-mode
resonator to feed a patch for enhancing the bandwidth of the
patch antenna is proposed. However, most of these works
focus on single-band antenna. In this paper, the concept in
[20] is further developed to design DBDP antenna with a low
frequency ratio. By employing resonator as the feed of the
antenna, not only the RF frontend can be simplified, but also
the frequency responses of the antenna can be improved. In
this work, two nested patches are coupled and tuned to a dual-
mode stub-loaded resonator (SLR), generating two operation
bands simultaneously. The coupling strengths between the
SLR and the patches can be tuned by adjusting the position
and the
A Shared-Aperture Dual-Band Dual-Polarized
Filtering-Antenna-Array with Improved
Frequency Response
Chun-Xu Mao, Steven Gao, Member, IEEE, Yi Wang, Senior Member, IEEE, Qi Luo, Member, IEEE,
Qing-Xin Chu, Senior Member, IEEE
W
For Review Only
dimension of the coupling slot. In addition, the antenna
exhibits 2
nd
-order filtering characteristics at both bands with
improved bandwidths and out-of-band rejections. A proof-of-
concept 2 × 2 array is conceived, prototyped and measured.
Such antenna can also be extended to a massive array or use as
the feed of a reflector antenna.
This paper is organized as follows. Section II introduces the
configuration of the DBDP element and its equivalent circuit.
Section III details the design methodology. Section IV
illustrates the proposed DBDP array. Section V presents the
results and followed by the conclusion.
II. DBDP ANTENNA ELEMENT DESIGN
A. Configuration
Fig. 1 shows the configuration of the proposed dual-band
dual-polarized antenna element, which is composed of a stacked
structure. The low and high band radiation elements are printed
on the top- and bottom-layer of the upper substrate, respectively.
As shown in Fig. 1(a), the low band antenna is a perforated
patch with the perforation is large enough for allowing the
radiation of the high band square patch. The size of the square
patch is approximately a half-wavelength of the high band
operation, whereas the perimeter of the perforated patch is about
one wavelength of the low band operation. The two patches are
nested in order to be fed by a common feed. For each
polarization, a folded SLR is used to feed the two patches
through a single slot in the ground. To improve the isolations
and reduce the cross polarization, the two coupling slots for the
each polarization are placed perpendicularly with each other.
Fig. 1(b) shows the configuration of the feeding networks
printed on the bottom layer of the lower substrate. Because of its
flexibility in controlling the resonant frequencies, the dual-mode
SLR is used here so that it can be tuned with the two patches.
The two resonant modes of the SLR can be derived using the
odd- and even-mode method [20]-[22],
rrr
odd
LL
c
f
)2(*2
31
(1)
rrrr
even
LLL
c
f
)2/(*2
321
(2)
where c is the light velocity in the free space, ε
r
is the effective
dielectric constant, f
odd
and f
even
are the odd- and even-mode
resonant frequencies of the SLR, respectively. In this design, f
odd
is tuned with low-band perforated patch while f
even
is tuned with
the high-band square patch. Using the equation (1)-(2), the two
resonant frequencies can be controlled independently by tuning
the lengths of the resonator L
r1
, L
r2
and L
r3
. Since the two bands
of each polarization are fed using a single SLR in this design,
the number of the feeding structures as well as the input ports
can be reduced by a half.
The stacked configuration of the antenna is shown in Fig.
1(c). A Rohacell 51HF foam with a thickness of 2 mm is
inserted between the two substrates as a spacer to improve the
antenna impedance matching and bandwidths. RT/Duriod
4003C substrate with a relatively dielectric constant of 3.55 and
loss tangent of 0.0027 is used in the design. High Frequency
Simulation Software (HFSS 15) is employed to perform the
simulations and the optimized parameters are listed in Table I.
B. Topology and synthesis
The proposed dual-band element for each polarization can be
represented and synthesized using a coupled resonator-based
topology, as shown in Fig. 2. The two patches can be regarded
as two single-mode resonators or a dual-mode resonator with
corresponding resonant frequencies of f
1
and f
2
, respectively.
The odd- and even-mode of the SLR are also tuned to resonate at
TABLE I
PARAMETERS OF THE PROPOSEDANTENNA: (MM)
Lp1
Lp2
Lp3
Lp4
Ls1
Ls2
Ws
Lf1
Lf2
14.5
9
7.8
3
7.2
10
0.5
11.8
8.4
Wf
Lr1
Lr2
Lr3
Lr4
S
Hs
Hf
1.8
5.5
6
2
5.5
0.3
0.813
2
Rogers 4003
Rogers 4003
C-band patch
X-band patch
Ground plane
Coupling slots
Feeding networks
Lp1
Lp2
Lp3
Ls2
Ws
(a)
Lf1
Lr4
Lr2
Lr3
S
Wf
Lf2
Lr1
Port 2
Port 1
(b)
Rogers
4003
Foam
C-band patch
X-band patch
Feeding network
Ground
Hs
Hf
(c)
Fig. 1. Configuration of the proposed DBDP element: (a) exploded view,
(b) bottom view, (c) side view.
For Review Only
f
1
and f
2
. Then the SLR and the two patches are
electromagnetically coupled, forming two separate passbands
with 2
nd
-order filtering responses. The two passbands of each
polarization are excited simultaneously. The lines between the
resonators in the topology represent the coupling between the
resonators. As an example to implement a DBDP antenna with
low frequency ratio (less than 2), the frequency specifications of
the dual-band antenna are given as follows,
Low band: f
1
= 5.2 GHz, BW = 200 MHz
High band: f
2
= 10 GHz, BW = 500 MHz
The coupling coefficients and the external quality factors can
be derived,
Low band: m
1,2
= 0.031, Q
ext
= 45.1
High band: m
1,2
= 0.051, Q
ext
= 27.2
where m
i,j
is the coupling coefficient between the resonator i
and j.
III. METHODOLOGY
A. Coupling
In this integrated design, the two patches can be fed by the
SLR through electromagnetic coupling only if the two patches
are tuned with the dual modes of the SLR, respectively. Fig. 3
shows the simulated tangential magnetic-field distribution of an
unfolded SLR corresponding to the odd- and even-mode
resonance. As can be observed in Fig. 3(a), the SLR has the
strongest magnetic-field distribution at its central part (denoted
as point A) when the odd-mode is excited (low-band
operation). It is noted that the dashed rectangle indicates the
location of the coupling slot where the power is transferred to
the patches through a coupling. When the even-mode is excited
(high-band operation), there are two magnetic-field maxima, as
the point B and C indicated in Fig. 3(b). On the contrary, point
A becomes the minima. By positioning the slot between A and
C, the electromagnetic energy can be coupled to the two
patches simultaneously.
The coupling strengths at the two operation bands can be
tuned by adjusting the relative coupling positions (d) and the
dimension of the coupling slot [19]. According to experimental
studies, the approximate equations for evaluating coupling
strengths can be derived,
d
WsLsA
M
f
1
1
(3)
dWsLsAM
f
22
, A < d < C, (4)
where, M
f1
and M
f2
are the mutual couplings between the SLR
and the patch at f
1
and f
2
; Ls is the length of the coupling slot;
Ws is the width of the coupling slot; d is coupling position,
which is situated between the point A and C. A
1
and A
2
are the
corresponding constants. As can be observed, the mutual
couplings are proportional to the length and width of the
coupling slot. At the low-band (f
1
), M
f1
is inversely
proportional to the d. At the high-band (f
2
), M
f2
is proportional
to the
d.
By using full-wave simulations, the corresponding coupling
coefficients between the SLR and the patches can be evaluated
using the following expression [20]-[21],
22
22
ji
ij
ji
ff
M
ff
(5)
where f
i
and f
j
are the two resonant frequencies of the 2
nd
-order
coupled SLR-patch for each band operation. Fig. 4 presents the
coupling coefficients between the SLR and the two patches
corresponding to different lengths of the slot (denoted as Ls)
and the locations of the slot (denoted as d in Fig. 3). As can be
seen, the dimension of the slot has a similar influence on the
coupling strength at both bands. But the variation of the
coupling coefficients between the SLR and the square patch
(high band operation) is more significant than that of the
low-band. In contrast, changing the locations of the coupling
slot has a reverse effect on the two bands, as shown in Fig. 4(b).
This disparate relationship provides another degree of freedom
to obtain the coupling coefficients required for the two bands.
B. 2
nd
-order filtering characteristics
Owing to the dual-mode SLR is coupled and tuned with two
single-mode patches, 2
nd
-order filtering characteristics at the
both bands can be achieved. The advantages of this coupling
structure can be appreciated by the comparison with the case
that fed by a microstrip line. Fig. 5 shows the simulated
S-parameters of the coupled SLR-fed antenna and a counterpart
P1
P2
A
Odd mode
Ls
(a)
P1
P2
B
C
Even mode
A
d
(b)
Fig. 3. Tangential magnetic-field distribution on the surface of an unfolded
SLR: (a) the odd-mode resonance at 5.2 GHz and (b) the even-mode
resonance at 10 GHz.
SLR
Ant_2
Ant_1
f1
f2
f1
f2
Port
Fig. 2. The equivalent resonator-based topology of the proposed dual-band
filtering antenna at each polarization.
For Review Only
fed by microstrip. It should also be noted that the results of the
microstrip-fed antenna at the two bands are obtained separately
as different lengths of the microstrip are required to achieve
impedance matching at two bands. It is observed that the
proposed SLR-fed antenna exhibits a 2
nd
-order filtering
characteristics with improved bandwidth and out-of-band
rejection as compared with the counterpart of microstrip-fed.
For the microstrip-fed antenna, only one resonant frequency
can be seen in the low band and the fractional bandwidth
(FBW) is only 1.9%. However, when the patches are fed using
the resonator, two resonant frequencies are generated at both
bands, resulting in wider bandwidth (FBW = 4.6 %).
The current distribution at the two operation bands is shown
in Fig. 6. As can be observed, strong current flows along the
perforated patch at the low band operation. The current on the
inner square patch is very weak. However, when the antenna
works at high band, the square patch is excited with intensive
current distribution. In contrast, the current on the perforated
patch is very weak except from the circle area of the
perforation. The current distribution demonstrates that the
perforated patch operates at low band whereas the inner square
4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 5.6
-30
-25
-20
-15
-10
-5
0
S
11
(dB)
Frequency (GHz)
Ls1=8 mm
Ls1=7.5 mm
Ls1=7 mm
(a)
8.5 9.0 9.5 10.0 10.5 11.0 11.5
-30
-25
-20
-15
-10
-5
0
Ls1=8 mm
Ls1=7.5 mm
Ls1=7 mm
S
11
(dB)
Frequency (GHz)
(b)
Fig. 7. The variation of bandwidths with different lengths of the slot Ls1: (a)
low band, (b) high band.
(a)
(b)
Fig. 6. Current distribution at two bands: (a) 5.2 GHz, (b) 10 GHz.
7.5 8.0 8.5 9.0 9.5 10.0 10.5
0.02
0.04
0.06
0.08
0.10
0.12
0.14
0.16
Coupling coefficient
Ls (mm)
SLR-perforated patch
SLR-square patch
(a)
0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
0.01
0.02
0.03
0.04
0.05
0.06
SLR-perforated patch
SLR-square patch
Coupling coefficient
d (mm)
(b)
Fig. 4. Coupling coefficients between the SLR and the two patches with
different (a) length of slot Ls, (b) location of the slot d.
4.8 5.0 5.2 5.4 5.6 8.5 9.0 9.5 10.0 10.5 11.0
-40
-30
-20
-10
0
S
11
(dB)
Frequency (GHz)
microstrip feed
SLR feed proposed
P1
P2
Fig. 5. S-parameters comparison between the SLR-feed dual-band antenna
and a microstrip feed dual-band antenna.
For Review Only
patch works at the high band. The mutual coupling between the
two patches is considerable weak.
Another feature of the SLR-fed antenna is that the
bandwidths at the both operations can be controlled by
adjusting the coupling coefficient between the SLR and the two
patches. Using the results presented in Fig. 4, the coupling
coefficients can be adjusted by changing the dimensions of the
coupling slot. Thus, the bandwidths can be controlled
correspondingly. Fig. 7 shows the variation of the bandwidths
at both bands with different lengths of the coupling slot. As can
be observed, when the Ls1 is less than 7 mm, the two reflection
zeros in each band are merged together. When the length
increases from 7 to 8 mm, the two reflection zeros emerge and
the bandwidths are increased from 150 to 250 MHz (500 to 800
MHz) for the low band (high band) operations.
C. Harmonic suppression
The proposed antenna has another advantage of harmonic
suppression, as also shown in Fig. 5. For the microstrip-fed
antenna, a 2
nd
-order harmonic is emerged at around 9 GHz,
causing the channel interference for the high band operation.
However, when the patches are fed by the resonator, the
harmonic can be significantly suppressed with the return loss
suppressed from 15 to 2 dB. This harmonic suppression is
attributed to the integration of the filtering and the radiating
devices that only the frequencies that match the two resonant
frequencies of the SLR can be transmitted/received. To verify
the harmonic suppression performance, simulated current
distribution at 9 GHz is presented, as shown in Fig. 8. It can be
observed from Fig. 8(a), 2
nd
-order harmonic is stimulated on
the perforated patch with strong current flowing on the patch
when the antennas are fed by microstrip. However, when the
antennas are fed by SLR, the current on the perforated patch is
significantly reduced and the 2
nd
-order harmonic is suppressed,
as presented in Fig. 7(b). Such filtering characteristics is useful
for reducing the interferences from unwanted channels.
IV. DBDP FILTERING ARRAY
The presented DBDP filtering antenna element can be
extended to conceive antenna array. To verify this concept, a 2
× 2 array is designed, as shown in Fig. 9. The two bands for
each polarization are combined using an SLR and fed by a
power dividing network, resulting in the reduction of the
number of the coupling slots, feed networks and input ports.
Thus, only two feed networks are employed to feed the DBDP
array. This reduction is very useful for simplifying the
complexity of the designs of the dual-band antenna arrays with
dual polarizations. This improvement is very useful to design a
shared-aperture DBDP array with a low frequency ratio.
One of the problems to be solved in this integrated DBDP
array is that a wideband dividing network is required to evenly
feed the radiation elements. To overcome this problem, a 2-way
T-junction power divider with stepped quarter wavelength
transformer is conceived. Fig. 10 shows the configuration of the
4 5 6 7 8 9 10 11
-60
-50
-40
-30
-20
-10
0
S-parameters (dB)
Frequency (GHz)
S11
S21
S31
Port 1
Port 2
Port 3
1.2
0.9
0.5
0.5
4
4.5
Fig. 10. The simulated S-parameters of the stepped two-way power divider:
(unit: mm).
24
1.2
0.9
0.5
1.8
Port 1
Port 2
Fig. 9. The layout of the proposed 2 × 2 DBDP filtering antenna array: (unit:
mm).
(a)
(b)
Fig. 8. The current distributions at 9 GHz of the two dual-band antennas with
different feeds: (a) microstrip feed, (b) SLR feed.
