Abstract— A novel design concept of multi-mode filtering
antenna, which is realized by integrating a multi-mode resonator
and an antenna, has been applied to the design of dual-polarized
antenna arrays for achieving a compact size and high
performance in terms of broad bandwidth, high frequency
selectivity and out-of-band rejection. To verify the concept, a 2×2
array at C-band is designed and fabricated. The stub-loaded
resonator (SLR) is employed as the feed of the antenna. The
resonant characteristics of SLR and patch as well as the coupling
between them are presented. The method of designing the
integrated resonator-patch module is explained. This integrated
design not only removes the need for separated filters and
traditional 50 Ω interfaces, but also improves the frequency
response of the module. A comparison with the traditional patch
array has been made, showing that the proposed design has a
more compact size, wider bandwidth, better frequency selectivity
and out-of-band rejection. Such low-profile light weigh
broadband dual polarized arrays are useful for space-borne
synthetic aperture radar (SAR) and wireless communication
applications. The simulated and measured results agree well,
demonstrating a good performance in terms of impedance
bandwidth, frequency selectivity, isolation, radiation pattern and
antenna gain.
Index Terms— Antenna array, broadband, dual-polarization,
filtering antenna, synthetic aperture radar (SAR), stub-loaded
resonator (SLR).
I. INTRODUCTION
ECENTLY, space-borne synthetic aperture radar (SAR)
has become increasing important for earth observation due
to its capacity of operation under all weather conditions day and
night. As a key component of the space-borne SAR system, the
antenna is required to have compact size, light weight, low cost,
low profile, broad bandwidth and high frequency selectivity.
Such antennas are also required in terrestrial wireless
communication and satellite communications. In [1]-[3], dual-
This manuscript is submitted on March 20, 2015. This work is supported by
the project “DIFFERENT” funded by EC FP7 (grant no. 6069923). YW is
supported by UK EPSRC under Contract EP/M013529/1.
C. X. Mao, S. Gao and F. Qin are with the School of Engineering and Digital
Arts, University of Kent, UK. (e-mail: cm688@kent.ac.uk, s. gao@kent.ac.uk).
Y. Wang is with the Department of Engineering Science, University of
Greenwich, UK. (e-mail: yi.wang@greenwich.ac.uk).
Q. X. Chu with South China University of Technology, China (e-mail:
qxchu@scut.edu.cn).
Ινπυτ
Φιλτερ
Ματχηινγ
νετωορκ
Ραδιατορ
Αντεννα
Ινπυτ
Μυλτι−mοδε
Ρεσονατορ
Ραδιατορ
Ιντεγρατεδ φιλτερ−αντεννα
Ινπυτ
Ρεσονατορ Ρεσονατορ Ραδιατορ
Ιντεγρατεδ φιλτερ−αντεννα
(α)
(β)
(χ)
Fig. 1 The RF front-end subsystem: (a) traditional architecture, (b) single-mode
resonator feeding, (c) proposed dual-mode resonator feeding.
band or triple-band shared aperture arrays were proposed to
reduce the volume and weight of SAR system. Broad antennas
are often realized by adopting stacked parasitic patches with a
large air gap between them [1]-[10]. An H-shaped slot in the
ground plane can also increase the bandwidth at the expense of
higher backward radiation [11], [12]. An L-probe feeding has
also been used to compensate the inductance of the patch, so as
to improve its bandwidth [13]. These methods can be employed
to design broadband antennas with a bandwidth over 20 % at
the expense of a large thickness and complex structure of the
antenna.
To improve the signal quality and reduce the interferences
from the out-of-bands, band-pass filters are required in the RF
front-end systems. Usually, the filter and the antenna are
cascaded, which not only increases the volume, but also
degrades the frequency response due to the mismatch and extra
insertion loss caused by the interconnections between them. To
reduce the size of RF front end and achieve high performance
of broad bandwidth and higher frequency selectivity, it is
necessary to employ seamless integration of the filter and the
antenna and eliminate the interface between them [14], [15].
Multi-Mode Resonator-Fed Dual Polarized
Antenna Array with Enhanced Bandwidth and
Selectivity
Chun-Xu Mao, Steven Gao, Member, IEEE, Yi Wang, Senior Member, IEEE, Fan Qin, Qing-Xin Chu,
Senior Member, IEEE
R
Z
1
, l
a
l
b
(a)
Z
in, odd
Z
in, even
Z
1
, l
a
/2
Z
1
, l
b
Z
1
, l
a
/2
(b) (c)
Fig.2 SLR and its equivalent circuit: (a) geometry, (b) odd-mode equivalent, (c)
even-mode equivalent.
In this paper, a novel design method is proposed by using the
stub loaded resonator (SLR) as the feed of the antenna. The
odd- and even-mode of the SLR and the patch as a resonator are
studied to realize a broadband antenna with a low profile. The
coupling between the SLR and the patch is investigated to
control the bandwidth. To verify this design concept, a 2×2
antenna array is designed, fabricated and measured for
space-borne SAR applications. To better illustrate the merits of
this novel integrated design, a traditional cascaded filter-
antenna subsystem is also built for comparison. The results
demonstrate that the integrated design has much improved
performance in terms of broadband impedance matching, high
frequency selectivity, ports isolation, antenna gains, cross
polarization discrimination (XPD) and radiation efficiency.
This paper is organized as follow. Section II introduces the
front-end systems evolution and proposed design methodology.
Section III presents antenna design of dual polarized SLR-fed
antenna element and array. Section IV highlights the
advantages of the integrated design over the traditional
cascaded design through comparison. Section V presents the
measured results followed by conclusion.
II. DESIGN METHODOLOGY
A. Evolution of the front-end
It is well known that the RF front-end of advanced wireless
systems are required to have compact size, low cost and
multiple functions. Fig. 1(a) presents the traditional
architecture of a RF front-end, where the filter and antenna are
designed separately and cascaded with 50 Ω interfaces and
matching networks. Fig. 1(b) shows an integrated filtering
antenna, where the separated matching network is eliminated
and the filter and the antenna are integrated seamlessly. The
antenna radiator serves as the last resonator of the filter and the
resonators feed into the radiator through coupling. The
single-mode resonators and the antenna radiator are synchrono-
Port 1
Port 2SLR
Patch
l
patch
l
b
Bottom layer
Top layer
l
a1
l
a2
L
slot
W
slot
Fig. 3 The configuration of a SLR-patch bandpass filter. The coupling slot is
cut in the ground plane, a layer between the top and bottom layer.
usly tuned to generate an operating band equivalent to a
three-pole filter.
To make the integrated filter-antenna system more compact,
a multi-mode resonator-fed antenna is proposed in this paper,
as shown in Fig. 1 (c). The multi-mode resonator is coupled
with the antenna, effectively generating a multi-mode antenna
with broad bandwidth and filtering performance.
B. Study of the SLR and patch
To demonstrate the design concept of Fig.1 (c), a stub loaded
resonator (SLR) is adopted. As shown in Fig. 2 (a), the SLR is a
dual-mode resonator with two independently controllable
resonant modes. Due to the symmetric of the SLR, it can be
analyzed using the odd- and even-mode method [16]. Fig. 2 (b)
(c) present the equivalent circuits of the odd-mode and
even-mode, respectively. When the odd-mode is excited, the
voltage at the middle of the SLR is zero and the SLR is shorted
there, as depicted in Fig. 2(b). When the even-mode is excited,
the symmetrical plane can be viewed as a magnetic wall,
equivalent to an open circuit, as depicted in Fig. 2(c). The input
impedance of the odd-mode and even-mode can be derived as
,1
tan ( / 2)
in odd a
Z jZ l
(1)
1
,
tan ( / 2 )
in even
ab
Z
Zj
ll
(2)
where Z
1
is the characteristic impedance of the horizontal
microstrip line in Fig. 2(a), and is the propagation constant.
When the input impedance Z
in
becomes infinite, the odd-mode
and even-mode resonate. Their resonant frequencies are
2
odd
a eff
c
f
l
(3)
2 ( / 2 )
even
a b eff
c
f
ll
(4)
4.0 4.5 5.0 5.5 6.0
-70
-60
-50
-40
-30
-20
-10
0
S
21
(dB)
Frequency (GHz)
l
patch
=15.2 mm
l
patch
=15.4 mm
l
patch
=15.6 mm
odd-mode, f
1
even-mode, f
2
f
0
Fig. 4 The change of the resonant frequencies of the structure in Fig. 3 with
l
patch.
Fig. 5 The change of the resonant frequencies of the structure in Fig. 3 with l
a1.
where, 坐
eff
is the effective dielectric constant, c is the speed of
light. It is observed from (3) and (4) that the odd-mode and
even-mode resonant frequencies can be controlled by tuning
the lengths of the stubs.
To examine the coupling between the SLR and the patch, a
two-port circuit composed of a SLR and a patch on different
layers is constructed, as shown in Fig. 3. Rogers 4003
substrates with a dielectric constant of 3.55 and loss tangent of
0.0027 is used in the design. The SLR is printed on the bottom
layer of the lower substrate with a thickness of 0.813 mm and
the patch is printed on the top layer of the upper substrate with a
thickness of 1.525 mm. The resonator in Fig. 2 is folded for
compactness, as characterized by the two sections l
a1
and l
a2
,
respectively in Fig. 3. The SLR is coupled with the patch
through a slot in the ground plane with the dimension of l
slot
=7
mm and w
slot
=0.3 mm. The dimension of the square patch l
patch
is about half wavelength at the operating frequency. By
adjusting the coupling between the SLR and the patch, a pass
band with three transmission modes can be obtained, as shown
in Fig. 4. It is observed that the odd- and even-mode
frequencies f
1
and f
2
of the SLR are located at both sides of the
patch resonant frequency f
0
. f
0
changes with the length of the
patch l
patch
.
4.0 4.5 5.0 5.5 6.0
-70
-60
-50
-40
-30
-20
-10
0
S
21
(dB)
Frequency (GHz)
l
b
=8 mm
l
b
=8.4 mm
l
b
=8.8 mm
odd-mode, f
1
even-mode, f
2
f
0
Fig. 6 The change of the resonant frequencies of the structure in Fig. 3 with l
b.
6.0 6.2 6.4 6.6 6.8 7.0
0.01
0.02
0.03
0.04
0.05
0.06
0.07
0.08
Coupling Coefficient
L6 (mm)
Fig. 7 The variation of the coupling coefficient with the length of the slot.
Fig. 5 shows that the odd- and even-mode resonant
frequencies vary with the total lengths of SLR l
a
. It is observed
that the influence on the odd-modes is more significant than
that of the even-mode. Fig. 6 shows the resonant modes vary
with the stub length l
b
. It is observed that the even-mode
resonant frequency f
2
shifts to lower frequency when l
b
increases, whereas the f
0
and f
1
keep unchanged. Therefore, the
three resonant frequencies can be controlled independently by
tuning the dimensions of the patch and SLR. As a result, the
bandwidth can be controlled.
The coupling strength between the SLR and the patch also
has a significant influence on the bandwidth of the filtering
antenna. The coupling can be controlled by the length and
width of the slot in the ground. The coupling coefficient can be
extracted using [17],
22
22
ji
ij
ji
ff
M
ff
(5)
where f
i
is the resonant frequency of the SLR and f
j
is the
resonant frequencies of the patch, respectively. Using full-wave
4.0 4.5 5.0 5.5 6.0
-70
-60
-50
-40
-30
-20
-10
0
S
21
(dB)
Frequency (GHz)
l
a1
=7 mm,
l
a2
=10 mm
l
a1
=8 mm
, l
a2
=10 mm
l
a1
=9 mm
, l
a2
=10 mm
odd-mode, f
1
even-mode, f
2
f
0
0.2 0.3 0.4 0.5 0.6 0.7 0.8
0.02
0.04
0.06
0.08
0.10
0.12
0.14
Coupling Coefficient
W1 (mm)
Fig. 8 The variation of the coupling coefficient with the width of the slot.
Λ1
Λ6
W1
Λ5
Λ3
Λ2
W2
D
Λ4
Η2
Η1
Πορτ 1
Πορτ 2
Fig. 9 The configuration of SLR-fed dual polarized antenna element.
simulation, f
i
and f
j
can be obtained and the coupling coefficient
can be calculated.
Fig. 7 shows that the simulated coupling coefficient between
the SLR and patch varies with the lengths of the slot. It is
observed that the coupling coefficient increases when the
length increases. Similarly, when the width of the slot increases,
a stronger coupling can also be observed, as depicted in Fig. 8.
In this design, L6 = 6.8 mm and W1 = 0.3 mm are chosen.
III. DUAL-POLARIZED ANTENNA ARRAY DESIGN
A. SLR-fed antenna element
Fig. 9 shows the configuration of one dual-polarization
integrated filtering-antenna element. The radiating element on
4.0 4.5 5.0 5.5 6.0
-30
-20
-10
0
S
11
(dB)
Frequency (GHz)
L6=6.0 mm
L6=6.4 mm
L6=6.8 mm
Fig. 10 The variation of the bandwidth with L6.
4.0 4.5 5.0 5.5 6.0
-30
-20
-10
0
S
11
(dB)
Frequency (GHz)
W1=0.2 mm
W1=0.3 mm
W1=0.4 mm
Fig. 11 The variation of the bandwidth with W1.
the top layer is fed by the SLR in the bottom layer via the slot in
the ground plane in the middle layer. The antenna and the
feeding network are printed on two substrates of Rogers 4003
with the thickness of 1.525 mm and 0.813 mm, respectively.
The two slots in the ground plane are placed perpendicular to
each other to reduce the coupling between the two
polarizations. Different from the traditional aperture coupled
patch antenna using the microstrip line, the patch in this design
is coupled to a resonator. The SLR in positioned in relative to
the slot in such a way as to obtain strong magnetic coupling
between them. In order to realize a wideband performance, the
odd- and even-mode of the SLR are placed at both sides of the
resonant frequency of the patch. The simulation were
performed using High Frequency Simulation Software (HFSS)
and the optimized parameters are presented in Table 1. It should
be noted that the thickness of the whole module is 2.338 mm
with two substrates, which is much lower than the previous
work in the references [1]-[13].
For the SLR-fed antenna, the bandwidth can be controlled by
Table 1: Parameters of the filtering antenna: (mm)
L1
L2
L3
L4
L5
L6
30
14
5.9
8.4
7.4
6.8
W1
W2
D
H1
H2
0.3
0.2
1.4
0.813
1.525
Πορτ 1
Πορτ 2
30mm
1mm
0.45mm
1.8mm
0.45mm
17.8mm
Fig. 12 The layout of the 2
2 SLR-fed dual polarization array.
v
(a) (b)
Fig. 13 The prototype of the 2×2 dual polarization SLR-fed array: (a) top layer,
(b) bottom layer.
tuning the dimension of the SLR or the coupling strength
between the SLR and the patch. Fig. 10 shows that the
bandwidth of the antenna vary with different lengths of the slot.
It is observed that the first and the second resonant frequencies
separate when L6 increases, resulting in a wider bandwidth.
Similarly, when the width W1 increases, the coupling strength
and bandwidth also increase, as shown in Fig. 11.
B. SLR-fed 2×2 dual-polarized array
Fig. 12 shows the layout of the 2×2 dual polarization antenna
array fed by SLRs for the SAR applications. The array works in
C-band with the central frequency of 5.2 GHz. The spacing
between the elements is 30 mm (0.53
0
). For X-direction
polarization (excitation from port 1), a four-way T-shaped
power divider is used to feed the SLRs. For Y-direction
polarization (excitation from port 2), an out-of-phase T-shaped
power divider is adopted to feed the SLRs. The 180
o
phase
difference is achieved by extending one branch of transmission
lines by half of a guided wavelength (i.e. 17.8 mm). The
out-of-phase power divider here is used to improve the cross
polarization discrimination (XPD) [18]. Quarter-wavelength
impedance transformer is used for impedance matching. The
size of the 2×2 array is 60 mm ×60 mm with a thickness of
2.338 mm. Fig. 13 shows the top and bottom layers of the
prototype array.
50 м ιντερφαχε
ΣΛΡ φιλτερ
4 ελεmεντ πατχη αρραψ
Ινπυτ πορτ
Fig. 14 Configuration of the conventional cascaded filter-antenna. Only one
polarization is implemented.
3.8 4.0 4.2 4.4 4.6 4.8 5.0 5.2 5.4 5.6 5.8 6.0 6.2
-45
-40
-35
-30
-25
-20
-15
-10
-5
0
S11 (dB)
Frequency (GHz)
SLR bandpass filter
traditonal patch array
cascaded filter-antenna
integrated filter-antenna
Fig. 15 The S11 comparison between four devices: the SLR filter, the parch
array, the cascaded filter-antenna and the integrated filter-antenna.
IV. COMPARISON WITH CONVENTIONAL DESIGN
To better illustrate the merits of the integrated filter-antenna
design, a traditional cascaded filter-antenna module is built and
compared, as shown in Fig. 14. Without loss of generality, only
one polarization is implemented. The bandpass filter and the
patch array are designed independently and then cascaded with
the 50 Ω interface. The bandpass filter is composed of the
dual-mode SLR. The four-element patch array is directly fed by
a T-shape four-way power divider via coupling slots. The input
impedance of the antenna array is set as 50 Ω. Firstly, it is
evident that the size of this cascaded structure is larger than the
integrated structure in Fig. 12.
Fig. 15 compares the impedance bandwidth of the SLR filter,
the standalone patch array, the cascaded filter-antenna and the
integrated filter-antenna. As can be seen, the traditional patch
array has a -10 dB bandwidth of only 150 MHz, mainly due to
the narrowband nature of the patch. Fig. 15 also shows the
standalone SLR filter has the 2
nd
order characteristics with two
poles and a bandwidth of 350 MHz. For the cascaded module,
the overall bandwidth is largely limited by the narrowband
component which is the patch array in the design. This is
evident shown from Fig. 15 that a bandwidth of only 170 MHz
for the cascaded filter-antenna is achieved. This is much
smaller than the integrated design, which exhibits a bandwidth
