Periodic array of complementary artificial
magnetic conductor metamaterials-based
multiband antennas for broadband wireless
transceivers
ISSN 1751-8725
Received on 25th January 2016
Revised on 19th May 2016
Accepted on 9th June 2016
doi: 10.1049/iet-map.2016.0069
www.ietdl.org
Mohammad Alibakhshi-Kenari
1
, Mohammad Naser-Moghadasi
1
✉
, Ramzan Ali Sadeghzadeh
2
,
Bal S. Virdee
3
, Ernesto Limiti
4
1
Faculty of Engineering, Science and Research Branch, Islamic Azad University, Tehran, Iran
2
Faculty of Electrical Engineering, K.N. Toosi University of Technology, Tehran, Iran
3
Center for Communications Technology, Faculty of Life Sciences and Computing, London Metropolitan University, London N7 8DB, UK
4
Dipartimento di Ingegneria Elettronica, Università degli Studi di Roma Tor Vergata, Via del Politecnico 1, 00133 Roma, Italy
✉ E-mail: mn.moghaddasi@srbiau.ac.ir
Abstract: This study presents the empirical results of a low-profile light-weight antenna based on a periodic array of the
complementary artificial m agnetic conductor metamaterial structure, which is realised by l oading the antenna with E-
shaped slits and inductive microstrip lines grounded using metallic via-holes. The finalised prototype antenna operates
over a broadband of 0.41–4.1 GHz, which corresponds to a f ractional bandwidth of 165.84%, and has dimensions of
40 × 35 × 1.6 mm
3
or 0.054
l
0
× 0.047
l
0
× 0.0021
l
0
, where
l
0
is free-space wavelength at operating frequency of 410 MHz.
The fina lised antenna has a peak gain and radiation efficiency of 4.45 dBi and 85.8%, respective ly, at 2.76 GHz. At the
lower operating frequency of 410 MHz, the gain and radiation efficiency are 1.05 dBi and 32.5%, respectively, which is
normally highly challenging to realise with very small antennas. The planar nature of antenna enables eas y integration
with wireless transceivers.
1 Introduction
Rapid development of wireless communication systems is bringing
about a wave of new wireless devices and systems to meet the
demands of multimedia applications [1, 2]. Multi-frequency and
multi-mode devices such as cellular phones, wireless local area
networks and wireless personal area networks place several
demands on the antennas [3, 4]. Primarily, the antennas need to
have high gain, small physical size, broad bandwidth, versatility,
embedded installation and so on. In particular, the impedance
bandwidth, radiation patterns, gains and efficiencies are the most
important factors that affect the application of antennas in
contemporary and future wireless communication systems [5].
Research into metamaterials based on periodic unit cells has
grown rapidly with the discovery of left-handed metamaterials [6,
7]. The transmission-line metamaterial (TL-MTM) technology has
been applied to various types of antennas, i.e. compact antennas
[8–12], leaky-wave antennas [13] and series-fed antenna arrays
[14]. In [7], the bandwidth and radiation characteristics of
monopole antennas, which are based on T- and F-shaped radiators,
are loaded with split-ring resonators of various sizes. In [9–12],
the antenna designs are based on composite right/left-handed
transmission-line unit cells implemented by engraving slits on the
radiating patch that is loaded with spiral inductors. This
technology brings enabling capabilities, in particular: (i) the ability
to strongly manipulate the propagation of electromagnetic (EM)
waves in confined small structure; and (ii) the freedom to precisely
and systematically determine a broad set of parameters including
the bandwidth, gain, efficiency and physical size. Therefore,
TL-MTMs appear to be a suitable candidate for developing
electrically small antennas for multiband applications.
In this paper, a new artificial magnetic conductor (AMC) structure
has been proposed for a low-profile antenna that offers enhanced
gain performance, reduced back-lobe level radiation, wideband
operation and small physical footprint. The ground plane of the
antenna acts like a reflector for the radiation impinging on its
surface; however, the phase of radiation impinging on conductor’s
surface is reversed which can interfere destructively with the
radiated waves from the antenna. This can significantly reduce the
antenna’s radiation efficiency. However, if the reflected radiation is
in-phase (i.e. 0°) or with phase change in the range ±90°, then
there will be a constructive interference of reflected wave with the
radiated wave. This feature is exploited in this paper to improve
the radiation characteristics of the antenna over its operating
frequency range where the incoming radiations are reflected at
angle between ±90°. To accomplish this, the ground plane of the
proposed antenna is loaded with periodic array of E-shaped
complementary AMC (CAMC) and grounded inductive lines to
realise metamaterial properties. The work is organised as follows.
Section 2 describes the design process of implementing the
proposed antennas and presents the simulated and measured results
of the prototype antennas. Finally, the work is concluded in
Section 3.
2 AMC surfaces
In the analysis, the cross-section of the antenna structure in Fig. 1 is
considered. A cavity is formed by the perfect electric conductor
(PEC) ground plane and a partially reflective surface (PRS) placed
at a distance (h) as described in [15]. The PRS is assumed to be a
homogeneous surface in the analysis. The antenna function can be
described by following the paths of the waves undergoing multiple
reflections inside the cavity. Phase shifts are introduced by the
path length, the PEC and the reflection coefficient of the PRS.
Following the paths of the direct and the reflected waves and
taking into account the various phase shifts introduced to them,
the resonance condition of the cavity can be easily derived. The
PEC introduces a phase shift of π. The resonance condition can be
easily derived by imposing the phase difference of the transmitted
waves to be zero. The PRS introduces a phase shift equal to the
phase of its transmission coefficient,
f
T
.If
f
2
−
f
1
is the phase
IET Microwaves, Antennas & Propagation
Research Article
IET Microw. Antennas Propag., 2016, Vol. 10, Iss. 15, pp. 1682–1691
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difference between direct and reflected waves, the resonance
condition is written as follows:
f
2
−
f
1
= 2
f
T
−
2
p
l
2h −
p
= 2N
p
, N = 0, 1, 2 ... (1)
This resonant cavity behaves as a perfect magnetic conductor (at
normal incidence) since it reflects normal incident waves with zero
phase shift. Consequently, placing a simple point source in close
proximity to the PRS would result in constructive interference
between direct and reflected waves at the cavity resonance.
2.1 Antenna #1
The geometry of Antenna #1 structure consists of two square
radiation patches of different dimensions that are interconnected to
each other with a microstrip line. Two inverted L-shaped inductive
lines are attached between the interconnecting line and the input
feed-line plane, and are grounded at the bends using metallic
via-holes to implement the shunt left-handed inductance that
reflects the incident waves with reflection phase of near zero
degrees. This proposed structure helps to enhance the antenna’s
impedance bandwidth and also determines the antenna’s
unidirectional radiation.
The E-shaped CAMC unit cells create a resonance cavity in the
antenna structure. As explained above, the distance between PEC
and PRS must be such that the reflected waves through the PRS
into space have equal phases in the normal direction. The
resonance condition of the cavity is determined by (1). The
E-shaped CAMC unit cells essentially concentrate the EM fields
and currents near the antenna structure to effectively prevent the
fields from spreading along the antenna’s ground plane and
therefore contribute towards unwanted coupling. This technique
should allow implementation of small antennas with minimal
mutual coupling which is important to decorrelate multipath
channels in small cellular systems.
Disposed on either side of the two square radiating patches are
rectangular conductors that are loaded with periodic array of
E-shaped slits, as shown in Fig. 2a. The slits act as CAMC unit
cells exhibiting series left-handed capacitances [9–12]. Details of
this technique are described in [16, 17] where an AMC surface is
created using square and rectangular loop slits. The AMC surfaces
exhibit high surface impedance, due to which the magnetic field
tangential to the surface vanishes. Hence, the structure reflects
incident waves with zero or near zero phase shift. The proposed
Fig. 1 Antenna cross-section where a resonant cavity is created by PEC
and PRS
Fig. 2 Perspective view and fabricated prototype of Antenna #1
a Perspective view of Antenna #1
b Fabricated prototype of Antenna #1. Dimensions are given in Table 3
Fig. 3 Simulated and measured reflection coefficient and VSWR of Antenna
#1
a Simul ated (solid line) and measured (dashed line) reflection-coefficient response
b Simulated (circular line) and measured (square line) voltage standing wav e ratio response
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antenna is excited through an SMA connector located at the lower
end of the smaller square patch. The CAMCs unit cells produce
fringing fi elds that make the effective dimensions of the patch
greater than its physical dimensions. The size and number of
CAMC unit cells were determined through optimisation using
three-dimensional (3D) full-wave EM simulator so that CAMC
reflects incoming radiations at zero degrees. The goal here was to
design and implement the antenna within an area of 40 mm
2
.It
was necessary to optimise the antenna’s performance in terms of
its impedance bandwidth and radiation characteristics. The antenna
is constructed on Rogers RT/Duroid 5880 substrate with a
dielectric permittivity (ɛ
r
) and loss tangent (tanδ) of 2.2 and
0.0009, respectively. The substrate has a height (h) of 1.6 mm.
The dimensions of Antenna #1 are: 27 × 35 × 1.6 mm
3
or
0.036
l
0
× 0.046
l
0
× 0.0021
l
0
in terms of free-space wavelength at
400 MHz.
Although the single E-shaped slit is a narrowband resonant
structure, the extension in the bandwidth is achieved by
electromagnetically coupling multiple E-shaped slits, as shown in
Fig. 2, which is analogous to cascading together several identical
narrowband resonators [9–11]. The simulated and measured
reflection coefficient and Voltage Standing Wave Ration (VSWR)
response in Fig. 3 confirm that the antenna exhibits an impedance
bandwidth of 1.9 GHz from 0.4 to 2.3 GHz for S
11
< −10 dB, and
its measured VSWR < 2 between 0.4 and 3.0 GHz. The antenna
resonates in its operating range at four distinct frequencies, i.e.
700, 1250, 1820 and 2080 MHz. The proposed antenna operates
over ultra high frequency (UHF), L- and S-bands. Simulations and
measurements were carried out using High Frequency Simulator
Structure (HFSS) software and Agilent N5224A vector network
analyser, respectively. The proposed antennas were fabricated
using standard manufacturing techniques.
Agilent N5224A vector network analyser used to characterise the
antenna was calibrated with a standard short-open-load thru
calibration procedure. The simulated and measured results
presented agree well with each other, showing the accuracy of the
model is valid. The radiation pattern of the antenna was measured
in a standard anechoic chamber. The antenna gain was measured
using the comparative method that involves measuring the signal
received by a reference antenna and by the antenna under test
(AUT), and determining the relative difference in the gain of both
antennas when both the reference antenna and AUT are working
in the received mode [18]. With this information, the gain of the
test antenna is determined.
The measured radiation characteristics of the antenna at the
different operating frequencies are shown in Fig. 4. The antenna
radiates unidirectionally in both E- and H-planes. The
cross-polarisation is less than −20 dB for both planes, and the
radiation characteristics essentially remain stable over the
antenna’s operating frequency range. The simulated and measured
gain and radiation efficiency of Antenna #1 is shown in Fig. 5.It
is evident that the measured gain and effi ciency have a peak of
Fig. 4 Measured 2D radiation patterns of Antenna #1 at various operating frequencies of 0.4, 1.25, 2.08 and 2.3 GHz
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2.21 dBi and 51.4%, respectively, at the fourth resonant mode of
2.08 GHz. Salient features of Antenna #1 are tabulated in Table 1.
2.2 Antenna #2
Besides the requirement of compact size, the proposed antenna needs
to possess wide bandwidth and good radiation characteristics such as
gain and efficiency performance. These characteristic features are
conventionally achieved by increasing the effective cross-sectional
area of antenna. The impedance bandwidth and radiation
characteristics of the proposed antenna were improved by simply
employing five additional E-shaped CAMC unit cells that are
located above the larger patch, as shown in Fig. 6, with the
advantage of not affecting the radiating patch’s physical
dimensions. The impedance bandwidth and radiation
characteristics of Antenna #2 are shown in Figs. 7–9. Antenna #2
has a size of 33 × 35 × 1.6 mm
3
or 0.047
l
0
× 0.050
l
0
× 0.0022
l
0
,
where the free-space wavelength is 430 MHz.
The measured impedance bandwidth in Fig. 7 extends from 0.43
to 2.95 GHz for S
11
< −10 dB, which corresponds to 149.11%; thus
covering UHF, L- and S-bands. The antenna has VSWR < 2 between
0.4 and 3.0 GHz. The antenna resonates in its operating range at the
following frequencies: 0.75, 1, 1.25, 1.75, 2.05, 2.30 and 2.65 GHz.
The antenna radiates unidirectionally in both E- and H-planes, as
shown in Fig. 8, and its radiation characteristics are stable over
the antenna’s operating frequency range. Fig. 9 shows that the
antenna’s simulated and measured gain and radiation efficiency
performance have a peak value of 3.12 dBi and 52.7%,
respectively, at the sixth resonant mode of 2.30 GHz. Salient
features of the Antenna #2 are tabulated in Table 2.
2.3 Antenna #3
The next iteration is Antenna #3, which includes five additional
E-shaped CAMCs unit cells located above the larger square patch.
Table 1 Radiation characteristics of Antenna #1
Frequency, GHz f
start
:
0.4
f
r1
:
0.7
f
r2
:
1.25
f
r3
:
1.82
f
r4
: 2.08
(max)
f
end
:
2.3
simulated gain, dBi 0.65 0.92 1.53 2.0 2.48 2.25
efficiency,
%
30.3 34.7 40.1 46.4 54.7 51.5
measured gain, dBi 0.5 0.73 1.2 1.88 2.21 2.0
efficiency,
%
28.7 31.1 37.3 44.9 51.4 49.6
Fig. 5 Simulated (solid line) and measured (dashed line) gain and
efficiency response of Antenna #1
Fig. 6 Fabricated prototype of Antenna #2. Dimensions are given in
Table 3
Fig. 7 Simulated and measured reflection coefficients and VSWR of
Antenna #2
a Simul ated (solid line) and measured (dashed line) reflection-coefficient response
b Simulated (circular line) and measured (square line) voltage standing wav e ratio response
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Fig. 8 Measured 2D radiation patterns of Antenna #2 at various operating frequencies of
a 430 MHz
b 1 GHz
c 2.3 GHz
d 2.95 GHz
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