4362 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 54, NO. 12, DECEMBER 2006
Monolithic Broadband Gilbert Micromixer
With an Integrated Marchand Balun
Using Standard Silicon IC Process
Sheng-Che Tseng, Student Member, IEEE, Chinchun Meng, Member, IEEE, Chia-Hung Chang, Chih-Kai Wu, and
Guo-Wei Huang
Abstract—A single-ended wideband downconversion Gilbert
micromixer is demonstrated in this paper using 0.35-
m SiGe
BiCMOS technology. A transimpedance amplifier with resistive
feedback is utilized in the IF stage while a broadband Marchand
balun is employed to generate wideband differential local oscil-
lator signals. The planar Marchand balun topology employed in
this paper can generate truly balanced signals even in the presence
of the lossy low-resistivity (
10
cm) silicon substrate. A
systematic approach to measure the frequency response of each
individual stage in a Gilbert mixer is developed in this paper. This
single-ended wideband mixer has the conversion gain of 15 dB,
IP
1dB
of 19 dBm,
IIP
3
of 7 dBm, and the noise figure of
13 dB. The mixer works from 3.5 to 14.5 GHz.
Index Terms—Downconverter, Marchand balun, micromixer,
SiGe BiCMOS, silicon substrate, transimpedance amplifier (TIA),
wideband.
I. INTRODUCTION
T
HE ERA of the wireless applications with high data-rate
transmission and multiple functions is coming, e.g., the
IEEE 802.11a/b/g combo system [1], ultra-wideband (UWB)
system [2], and WiMAX system [3]. The range of carrier fre-
quencies and their bandwidth constantly increase. The obliga-
tion of the complicated data processing belongs to the baseband
design, while the RF integrated circuit (IC) design takes respon-
sibility for the wide range frequency and broad bandwidth oper-
ation. Nevertheless, the design of the high-frequency and wide-
band RF circuits is a big challenge in the overall solution im-
plementation. For an active mixer, the transistors have natural
instinct to perform wide range and broad bandwidth frequency
translation. Due to the input/output matching networks, narrow-
band passive components, and loading effects, the mixer’s wide-
band ability is restricted.
Manuscript received March 31, 2006; revised July 25, 2006. This work was
supported by the National Science Council of Taiwan, R.O.C., under Contract
NSC 95-2752-E-009-001-PAE and Contract NSC 95-2221-E-009-043-MY3, by
the Ministry of Economic Affairs of Taiwan under Contract 94-EC-17-A-05-S1-
020, by the Ministry of Education Aim for Top University Program under Con-
tract 95W803, and by the National Chip Implementation Center.
S.-C. Tseng, C. Meng, and C.-H. Chang are with the Department of Commu-
nication Engineering, National Chiao Tung University, Hsinchu 300, Taiwan,
R.O.C. (e-mail: ccmeng@mail.nctu.edu.tw).
C.-K. Wu was with the Department of Communication Engineering, National
Chiao Tung University, Hsinchu 300, Taiwan, R.O.C. He is now with Agilent
Technologies, Tao-Yuan 324, Taiwan, R.O.C.
G.-W. Huang is with National Nano Device Laboratories, Hsinchu, 300
Taiwan, R.O.C.
Digital Object Identifier 10.1109/TMTT.2006.884690
For the wideband circuit design, the wideband matching of
the input/output ports is a significant issue. The common imple-
mented active mixer is a Gilbert mixer using the emitter-coupled
differential input stage. Owing to the high input impedance
of the common-emitter-configured transistors, the reactive or
resistive matching is needed at the input port. For the reac-
tive matching, the matching bandwidth relates to the orders
of the passive matching network. Increasing the order of the
matching network can expand the operation bandwidth, but also
takes more area. Although the resistive matching can perform
wideband matching, it also introduces loss. The variant of the
Gilbert mixer, the so-called micromixer, which is defined as
a microwave mixer in [4], has the properties of the wideband
input matching and single-ended input. Those properties facil-
itate the realization of the wideband and single-ended mixer.
In this paper, the input stage of the mixer is made up of the
micromixer.
For the balanced mixers, the Gilbert switch quad demands
differential local oscillator (LO) signals. It is cumbersome to
use an off-chip balun for the wideband balanced LO signal gen-
eration because the differential signals experience the different
delay paths on the circuit board, especially at high frequen-
cies. Hence, a single-to-differential LO balun is integrated in
the IC process to form a single-ended mixer. Since it is difficult
to achieve truly differential signals with equal magnitude and
opposite phase by an active balun in addition to more power
consumption at high frequencies, a passive balun is taken into
consideration. The Marchand balun is a very wideband passive
balun and is popularly used for broadband applications such as
a double-balanced diode mixer [5] and a frequency doubler [6].
However, most Marchand baluns are realized on a semi-insu-
lating or high-resistivity substrate. The proper Marchand balun
topology suitable for a standard silicon IC process is identified
in this paper to maintain the truly balanced signals regardless of
the substrate loss.
High impedance resistors or active pMOS loads are usually
employed to obtain high conversion gain. In addition, the pMOS
current mirror is used to effectively combine the differential
IF output current signals of the mixer and establish a single-
ended output. However, the high impedance causes a low-fre-
quency pole at the output stage, which slows down the IF re-
sponse. The transimpedance amplifier (TIA) with resistive feed-
back is, hence, utilized at the output stage to reduce the output
impedance and extend the bandwidth in this paper [7], [8].
A single-ended wideband Gilbert downconverter is fabricated
in the 0.35-
m SiGe BiCMOS technology and demonstrated in
0018-9480/$20.00 © 2006 IEEE
TSENG et al.: MONOLITHIC BROADBAND GILBERT MICROMIXER WITH INTEGRATED MARCHAND BALUN 4363
Fig. 1. Planar Marchand balun with
=
4
coupled lines.
this paper. It is composed of a micromixer, an integrated LO
Marchand balun, and a TIA output amplifier. In this paper, a
technique to measure the RF, LO, and IF stages of a Gilbert
mixer is developed. This mixer has 15-dB conversion gain,
13-dB noise figure, and 400-MHz IF bandwidth and works
from 3.5 to 14.5 GHz.
The Marchand balun design concept and the measured re-
sults of a monolithic planar Marchand balun are represented in
Section II, and Section III depicts the entire circuits of the mi-
cromixer with an integrated Marchand balun. Section IV then
shows the experimented results, including the performances of
an individual micromixer and an overall micromixer with an in-
tegrated Marchand balun. Finally, Section V gives a conclusion
with a brief summary of the mixer’s performances.
II. A
NALYSIS AND IMPLEMENTATION OF THE PLANAR
MARCHAND BALUN ON SILICON IC PROCESS
A. Analysis
The Marchand balun, a very broadband passive balun, was
proposed in 1944 and has one unbalanced input and two bal-
anced outputs [9]. The compensated Marchand balun can per-
form impedance transformation from the balanced port to the
unbalanced port. The load at the balanced port is shunted with a
quarter-wavelength short stub and in series with a quarter-wave-
length open stub [10], [11]. Nevertheless, this type is not easily
realized in the IC process, especially in the silicon IC process,
and thus is not commonly used in ICs.
The planar Marchand balun is composed of two back-to-back
quarter-wavelength coupled lines, as shown in Fig. 1. Each cou-
pled line has four ports—input, direct, coupled, and isolated
ports. Two coupled ports of coupled lines are connected with
short ends; the direct ports are tied together. One of the input
ports is connected with an open end and the other is the unbal-
anced input of the Marchand balun, while the balanced outputs
of the Marchand balun are from the isolation ports. This con-
figuration is the most popular one, and other topologies of the
Marchand balun had been developed in [12].
The transmission and reflection properties of the Marchand
balun can be analyzed easily by the properties of the coupler
Fig. 2.
S
-parameter derivation of the planar Marchand balun with
=
4
coupled
lines. (a)
S
. (b)
S
.
and open and short terminals [13]. The quarter-wavelength cou-
pled line has the scattering parameters for the coupled and trans-
mitted ports
and , which are derived in Appendix. The re-
lation between
and of the coupled line is written as
no loss
with loss.
(1)
The short terminal results in an antiphase total reflection,
whereas the open terminal causes an in-phase total reflection.
When a signal inputs at port 1, one part of the input signal,
the solid line signal, shown in Fig. 2(a), couples to the short
terminal, then reflects totally in an antiphase fashion, and finally
transmits to port 2. This causes the
voltage wave
transmitting to port 2. The other part, the dotted line signal, is
analyzed more complicatedly, as shown in the following steps.
Step 1) The dotted line signal transmits to the open terminal
and reflects totally.
Step 2) Some reflected power directly transmits to the middle
and then couples to port 2; the rest power couples to
the short terminal
.
Step 3) A proportion of power reflected from the short
terminal
, couples to the open terminal, , and
then reflects totally. Finally, the reflected signal
progresses repeatedly from Step 2).
Consequently, the transmission coefficient form ports 1 to 2
caused by the dotted line signal is
. Therefore,
the total transmission coefficient from ports 1 to 2 is
(2)
With the same analysis approach, the total transmission coeffi-
cient form ports 1 to 3 is
(3)
as shown in Fig. 2(b). Based on the calculations of
and ,
this balun performs single-to-differential conversion perfectly,
regardless of the silicon substrate loss and metal loss, thanks
4364 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 54, NO. 12, DECEMBER 2006
Fig. 3.
S
-parameter derivation of the planar Marchand balun. The nodes (1–3) of the tree denote ports 1–3. The others nodes are denoted in Fig. 1.
Fig. 4. Magnitude of the transmission and reflection at the input port of the
planar Marchand balun with respect to coupling coefficients
k
for the cases of
no loss.
to the symmetric signal delivery, as shown in Fig. 2(a) and (b).
This procedure to figure out the scattering parameters can be
portrayed in the tree formation, as shown in Fig. 3. The tree
nodes
and , in Fig. 3, symbolize ports 1–3,
open end, and short terminal, respectively, as shown in Fig. 1.
As a result, the
-parameter matrix of the planar coupled-line
Marchand balun can be expressed as
(4)
Fig. 4 displays the magnitude of the transmission and reflec-
tion at the input port of two configurations with respect to dif-
ferent coupling coefficients
,asdefined in the Appendix with
the assumption of no loss. The coupling coefficient is designed
as the value of 1
3or 4.8 dB, for a good input matching,
i.e.,
, and the maximum transmission [14]. The optimal
Marchand balun is practically implemented on account of the
low optimal coupling coefficient.
B. Implementation
The coupled-line Marchand balun can be realized by Lange
couplers [15], [16], broadside coupled lines [12], [17], [18], and
spiral transmission lines [12], [17], [19]–[21]. In order to shrink
the size of the balun, an interleave transformer is employed as a
quarter-wavelength coupled line in our study, as shown in Fig. 1.
The transformer-type coupled lines, namely, spiral transmission
lines, can achieve the desired coupling coefficient. The coupled-
line Marchand balun with two short terminals and one open end
is applied and the two ac ground terminals tied together can
provide a dc bias for the mixer’s switch quad.
For the balanced mixers, the LO switch quad is driven by the
differential signals. A wideband single-to-differential Marchand
balun is demanded in order to offer differential LO signals and
to reserve the mixer wideband operation. Given that the input
impedance of the Gilbert cell
is not matched to the source
impedance
, the -parameters of Marchand balun are mod-
ified as shown in (5) at the bottom of the following page [14].
However, the balance of the two outputs is independent of the
coupling coefficient
and the load impedance . Even if the
load impedance of the Marchand balun is not matched, the out-
puts also have equal magnitude and opposite phase.
Most monolithic Marchand baluns are fabricated on the
semi-insulating GaAs substrate. A Marchand balun on the
high-resistivity (
4000 cm) silicon substrate had also been
demonstrated [22]. Recently, the Marchand balun was practiced
using standard silicon processing with a shielding ground plane
[23]. However, the shielding ground plane limits the even-mode
characteristic impedance and then reduces the balun bandwidth
[21]. The operating bandwidth of the Marchand balun increases
monotonically when the ratio of the even-mode character-
istic impedance to the odd-mode characteristic impedance of
the coupled line increases. A high even-mode characteristic
impedance is preferred for a wideband Marchand balun. Thus,
the high even-mode characteristic impedance of coupled lines
can be achieved in our Marchand balun topology to obtain wide
bandwidth. Besides, the higher effective dielectric constant for
the balun without the shielding ground plane is good for size
reduction.
In this paper, the planar Marchand balun, as shown in Fig. 5, is
implemented directly on the low-resistivity (
10 cm) silicon
TSENG et al.: MONOLITHIC BROADBAND GILBERT MICROMIXER WITH INTEGRATED MARCHAND BALUN 4365
Fig. 5. Die photograph of a monolithic Marchand balun. The connecting line is
approximately 180
m and is restricted by the GSGSG probe. From simulation,
the coupling coefficient of the coupled line is approximately 0.5 at 12 GHz.
(Color version available online at http://ieeexplore.ieee.org.)
substrate with the high even-mode characteristic impedance to
hold broadband operation. This Marchand balun is formed by
two-section transformer-type coupled lines and is designed at
the center frequency of 12 GHz. The size of Marchand balun
is approximately 660
m 250 m. It is very compact thanks
to the size advantage of the transformer-type coupled lines. The
coupled lines are made of the top metal with the thickness of
0.93
m, the spacing of 5 m, and the width of 5 m. The inter-
leave transformer has approximately 3 : 3 turns. The substrate
thickness is approximately 350
m and the distance between
the top metal and the substrate is approximately 6.2
m. From
simulation, the coupling coefficient of the coupled line is ap-
proximately 0.5 at 12 GHz. The experimental measured data,
IE3D simulation results, and the calculated data from (4) based
on IE3D simulated
and of the Marchand balun in Fig. 5
are displayed in Figs. 6 and 7. The delta plots of the phase
and amplitude errors are presented in Fig. 8. The magnitude
imbalance in output ports is approximately 2 dB. This magni-
tude imbalance results from the loss of the connecting line be-
tween two transformer-type coupled lines, as shown in Fig. 5.
The length of the connecting line is approximately 180
m. The
inevitable finite connecting line in the Marchand balun test pat-
tern is constrained by the ground–signal–ground–signal–ground
Fig. 6. Input return loss of the planar Marchand balun.
Fig. 7. Transmission coefficients of the planar Marchand balun.
(GSGSG) pad employed for the measurement purpose. The fi-
nite connecting line can be minimized in the final fabricated cir-
cuit. On the low-resistivity silicon substrate, the signal transmis-
sion of the balun is dominated by the first component, i.e., the
direct coupled term
of (2) and (3). However, the voltage
(5)
4366 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 54, NO. 12, DECEMBER 2006
Fig. 8. Phase error and magnitude error of the planar Marchand balun.
Fig. 9. Dissipated loss of the Marchand balun.
wave directly coupled to port 3 experiences the connecting-line
loss. Thus, the transmission magnitude
is lower than .
This phenomenon corresponds to the measured results. The out-
puts are more balanced in magnitude when the connecting line
is removed in the IE3D simulation, as shown in Fig. 7, but the
phase balance is almost unaffected by the connecting line. In
other words, the connecting line has high associated loss than
phase delay. The dissipated loss of a Marchand balun is defined
as
Loss
(6)
and is approximately 6 dB, as shown in Fig. 9. In our study, the
Gilbert mixer with the integrated Marchand balun has a short
connecting line to provide balanced outputs. The usable band-
width is more than 10 GHz.
The mixer conversion gain is insensitive to the LO power
provided that the phase is balanced and the LO power is large
enough to commutate the RF current. The reason will explained
by the measured results in Section V. The magnitude imbalance
resulting from the small connecting line loss is, hence, not a
matter of mixer’s operation. This balun is appropriately utilized
as a single-to-differential balun at the LO port in this mixer even
though the magnitude imbalance occurs.
Fig. 10. Schematic of the micromixer with an LO Marchand balun and a TIA
output buffer.
III. CIRCUIT
DESIGN
The entire schematic of the single-ended wideband downcon-
verter is shown in Fig. 10. This downconverter is formed by
the micromixer, the Marchand balun, and the TIA output buffer.
Each element has the broadband property.
The micromixer can be considered as the combination of two
single-balanced mixers. One mixer is formed by the common-
emitter-configured RF amplifier
; the other is composed of
the common-base-configured RF amplifier
. The LO switch
quad is made up of the transistors
, , and . The cur-
rent mirror pair
and provides the balance dc currents in
the RF input stage and then these two RF amplifiers have equal
magnitude and opposite phase transconductance gain to obtain
good mixer balance. Moreover, the diode-type transistor
re-
duces the input impedance of
and enhances the speed of the
common-emitter-configured input stage. The input impedance
is controlled by the transistors
and and the resistors
and . It is easy to achieve wideband matching so this mi-
cromixer can act as a wideband mixer [24].
To establish a single-ended output, the pMOS current mirror
is applied to combine the differential output current signals of
the mixer. Furthermore, a TIA amplifier is used in the output
stage of this mixer. The frequency response of the input stage
is dominated by the common-emitter-configured transistor
.
As shown in Fig. 11, in the critical path, the RF input stage is
viewed as a transconductance amplifier (TCA), the IF output
stage is a TIA, while the LO switch quad is inserted in the
middle and performs the frequency translation. The topology
is very similar to the well-known Cherry–Hooper amplifier—a
TCA stage in cascade with a TIA stage [7]. The LO current
commutation quad Gilbert mixer cell is used to switch the con-
necting current between the TCA and TIA stages. Thus, the con-
version gain and frequency response can be analyzed as a TCA
for the RF stage and a TIA for the IF stage. The TIA output
buffer employs a resistive feedback to enlarge the output band-
width. In addition, a Darlington pair is also utilized to enhance
the speed of transistors. Therefore, this output stage of the mixer
